High frequency switches are important to the operation of many types of modern systems, such as a radar, a phase array, and various types of wireless communications systems such as a mobile phone, a personal digital assistant (PDA), a global positioning system (GPS) receiver, and/or the like. Radio frequency (RF) switches also are essential elements of various phase shifters by adding/reducing phase delay when used in conjunction with transmission line segments. A traditional Silicon (Si) p-i-n diode-based RF switch leads to additional direct current (DC) voltage loss since the diode consumes power. Further, Silicon p-i-n diodes are difficult to adapt to applications that require both high continuous wave (CW) power handling and high switching rates. In particular, since RF energy is dissipated in the p-i-n diode during its relatively long switching transient, the “hot switching” power rating of a p-i-n switch is typically much lower than the steady-state value, even at low switching rates. Additionally, the p-i-n switches require driver circuits that tend to be complex, bulky, and expensive. As a result, a need exists to design switches based on active devices.
To date, most active device approaches have used Gallium Arsenic (GaAs)-based devices. For example, a GaAs-based field effect transistor (FET) can provide low-loss, high-frequency switching due to high electron mobilities in the FET. Further, the GaAs-based FET can be monolithically integrated with other GaAs microwave components, such as amplifiers, passive radio link control (RLC) components, and/or the like. However, GaAs-based switches cannot readily accommodate the high switching power required in communication and/or radar systems due to their relatively low peak currents. For higher power operation, the GaAs-based switches can be stacked, which complicates the driving circuits, reduces the bandwidth, and increases loss.
Other microwave device approaches have analyzed an AlGaN/GaN heterostructure field effect transistor (HFET). This device includes a high sheet carrier density (e.g., greater than 1.5×1013 cm−2, an order of magnitude higher than that for a GaAs/AlGaAs heterostructure) and a high room temperature mobility (e.g., greater than 2,000 cm2/Vs). The high sheet density and mobility of the two-dimensional electron gas (2DEG) electrons results in a very low sheet channel resistance, e.g., below 300Ω/square, and in record high saturation currents, e.g., well in excess of 1 A/mm.
Additionally, a GaN/AlGaN metal-oxide-semiconductor HFET (MOSHFET) has been shown to combine the advantages of the classical metal-oxide-semiconductor structure (e.g., a low gate leakage current) with that of the AlGaN/GaN heterointerface (e.g., high density, high mobility 2DEG channel). To this extent, the MOSHFET has extremely low gate currents with excellent gate control, which enables it to operate at a positive gate bias. As a result, the maximum saturation current achievable in a MOSHFET is nearly double that of a regular GaAs-bases HFET device. Additionally, the gate capacitance of the MOSHFET is lower due to a larger gate-to-channel separation, and the breakdown voltage is approximately the same or higher than that of an AlGaN/GaN HFET (e.g., as high as 500 Volts). These features indicate that the MOSHFET may be effective for use in high power, low loss RF switches, phase shifters, and attenuators. A similar device, commonly referred to as a MISHFET, includes Si3N4 instead of an oxide, such as SiO2, and also has been shown to provide a high performance insulated gate HFET. Both the MOSHFET and MISHFET operation characteristics are based on a high quality of the interface between the deposited SiO2/Si3N4 layer and the AlGaN barrier layer.
FIG. 1A shows an illustrative prior art MOSHFET. Similar to a regular AlGaN/Gan HFET, the built-in channel for the MOSHFET is formed by a high-density 2DEG at the AlGaN/GaN interface. However, in contrast to the regular HFET, the gate metal is isolated from the AlGaN barrier layer by a thin SiO2 film. As a result, the MOSHFET gate provides similar operating characteristics as an MOS gate structure, rather than a Schottky barrier gate used in a regular HFET. Since a properly designed AlGaN barrier layer is fully depleted, the gate insulator in the MOSHFET consists of two sequential layers, the SiO2 film and the AlGaN epilayer. The double layer ensures an extremely low gate leakage current and allows for a large negative to positive gate voltage swing.
The suppression of the gate leakage current is one of the most important features of the MOSHFET. FIG. 1 B shows current-voltage characteristics for a 1 μm gate MOSHFET and HFET measured at a drain voltage that is sufficient to shift the operating point into the saturation regime and the gate bias dependence of the MOSHFET and HFET gate currents in the saturation regime. As illustrated, for the HFET, gate voltages in excess of +1.2 V result in an excessive leakage current, which limits the maximum drain current. However, for the MOSHFET, gate voltages as high as +10 V could be applied, which results in an approximately one hundred percent increase in the maximum drain current with respect to a zero gate bias. However, the gate leakage remains well below 1 nA/mm.
An HFET or a MOSHFET can be used for high-power RF frequency switching. For example, FIG. 2A shows an illustrative circuit that includes a FET connected into a transmission line in series as a variable resistor. In particular, the source and drain electrodes are connected to the line input and output, respectively, while the gate electrode is connected to a control voltage supply through a blocking resistor. FIG. 2B shows a more detailed equivalent circuit of the FET RF switch, which includes the variable (gate voltage controlled) channel resistance, parasitic device capacitances, and inductors associated with the bonding wires.
When the FET comprises a MOSHFET, in an “ON” state, the MOSHFET gate bias is zero or positive, the channel resistance is low, which ensures a low-loss input-output transmission. The transmission is nearly frequency independent since the drain-source capacitance is shunted by the channel resistance. A value of the channel resistance, RCh, can be estimated from the device parameters as RCh=2RC+RGS+RGD+RG(VG), where RGS and RGD are resistances of the source-gate and gate-drain openings, respectively, RC is a contact resistance, and RG(VG) is a voltage dependent resistance of the channel under the gate. At a high positive gate bias, especially for a short gate device, RG(VG)<<RCh and RGS=RSH×LGS/W and RGD=RSH×LGD/W, where RSH=1/(qNSμn) is the layer sheet resistance, and LGS and LGD are the source-gate and gate-drain spacing, respectively.
For an illustrative MOSHFET, RSH≈400 LGS≈LGD≈1.5 μm and RC≈1 Ω/mm, which results in a RCh≈3Ω/mm. For a transmission line with a characteristic impedance, Z0=50Ω, and assuming RON<<Z0, the insertion loss, LIns, of the MOSHFET RF switch connected into the transmission line can be estimated as:
                                          L            Ins                    ⁡                      (            dB            )                          =                                            -              20                        ⁢                                                  ⁢            Log            ⁢                          1                              1                +                                                                            R                      ON                                        /                    2                                    ⁢                                      Z                    0                                                                                ≈                      0.087            ⁢                                          R                ON                            .                                                                      For a one millimeter wide MOSHFET with RON=RCh≈3Ω, LIns≈0.26 dB. This insertion loss is in strong agreement with FIG. 3, which shows experimental and simulated data for insertion loss as a measure of frequency.
When the MOSHFET is in the “OFF” state during operation, a gate bias is below a threshold voltage and a pinch-off channel current is as low as several nanoamperes. Correspondingly, a channel resistance increases and a switch transmission is determined by a source-drain capacitance. In this case, a switch isolation loss, Lls, can be estimated as
                    L        Is            ⁡              (        dB        )              =          20      ⁢                          ⁢      Log      ⁢              1                                        1            +                                                            Z                  DS                                /                2                            ⁢                              Z                0                                                                    ,where the impedance of the MOSHFET at the sub-threshold gate bias conditions, ZDS=1(jωCDS). As illustrated in FIG. 3, the simulated and experimental results of the MOSHFET switch isolation are in strong agreement.
Maximum switching power is an important parameter for an RF switch. To this extent, a III-N based MOSHFET switch provides a unique combination of vary high saturation currents and high breakdown voltage as compared to GaAs based devices. In an illustrative experiment, an RF signal generator (i.e., HP 8341 B by Agilent) was used in combination with a microwave power amplifier (i.e., HP 83020A by Agilent) to generate an RF signal with a maximum output power of approximately 28 dBm at 2 GHz. As shown in FIG. 4A, in the “ON” state, the saturation power of the 1 mm wide MOSHFET switch is well above the available output power level. To measure the maximum switching power, a MOSHFET with a total gate width of approximately 0.1 mm was used. FIG. 4B shows the experimental results when the MOSHFET with the smaller total gate width was used as an RF switch. As illustrated, when the MOSHFET has a zero volt gate bias, the insertion loss remains constant as the continuous wave (CW) input signal increases until the power level reaches approximately 25 dBm. At a higher positive gate bias (e.g., five volts), the insertion loss is less and the nonlinearity appears at a higher input power level, e.g., approximately 26.5 dBm. In contrast, an HFET exhibited both a higher insertion loss and a lower maximum input power level than that of the MOSHFET with the same gate bias.
The experimental results can be explained by considering the factors limiting RF switching power. A maximum power, PMON, that can be switched by the active element in the “ON” state can be expressed as PM=0.5IDS2×RL, where IDS is the drain saturation current at a given gate bias, and RL is the load resistance (normally 50Ω). For the 0.1 mm wide MOSHFET RF switch, the maximum load power, PM, should be: PM=0.5IDS2×RL=0.5×0.112×50=0.3W=24.8 dBm, which is very close to the measured maximum input power of 25 dBm. For the 1 mm wide MOSHFET with a drain saturation current of 1.1 Amps, the maximum switching power is PM≈30.2 W. This power is more than thirty times higher than that reported for GaAs RF switches. Additionally, the maximum switching power can be further increased by using a larger device periphery and/or positive voltage for biasing the MOSHFET gate into the “ON” state.
In the “OFF” state, the maximum switching power, PMOFF, is limited by the breakdown voltage of the MOSHFET, VBR, and can be calculated as PMOFF=VBR2/(2Z0). The breakdown voltage for the MOSHFET can be as high as 500 V. Using a moderate value of VBR≈200 V, the maximum power can be estimated as PMOFF≈400 W. AlGaN/GaN based MOSHFET devices for fast broadband high-power RF switching have been shown to have maximum switching powers in the range of 20-60 W/mm, well exceeding maximum switching powers achieved with GaAs technology.
Switch performance can be significantly improved by implementing monolithically integrated three-element π-type circuitry, and using sub-micron long gates to further decrease the channel resistance of the MOSHFETs. FIG. 5A shows a simplified π-type equivalent circuit of an RF switch Monolithic Microwave Integrated Circuit (MMIC). As shown, the series MOSHFET is marked as G1, and the two parallel MOSHFETs are marked as G2 and G3. In the “ON” state, the gate voltage applied to G1 is positive, and the transistor channel has low resistance. Additionally, the gate voltage applied to G2 and G3 is below the channel pinch-off voltage. In this case, the shunting effect of G2 and G3 is minor due to low parasitic capacitances between the source and drain electrodes of the MOSHFETs. As a result, the insertion loss of the π-type circuit should be close to that of the single element switch shown in FIG. 2A. In the “OFF” state, a voltage at the gate of G1 is negative and its channel is pinched off. Additionally, the voltage at G2 and G3 is positive. Low channel resistances of the MOSHFETs effectively shunt the circuit, thereby greatly increasing the isolation at high frequencies as compared to the single element switch.
FIG. 5B shows a charge-coupled device (CCD) image of an illustrative integrated RF switch, which was fabricated over an insulating silicon carbide (SiC) substrate using an epilayer design and MOSHFET fabrication technology. The sheet electron concentration in the 2DEG channel was approximately Ns ≈1.3×1013 cm−2, and the sheet resistance obtained by the on-wafer contactless resistivity mapping system was 290-310Ω. The 0.3 μm length gates were formed using e-beam lithography. The contact pad's layout was optimized to provide a 50Ω characteristic impedance. The MOSHFETs had a threshold voltage of VTH≈8 V, a maximum channel current IP≈1.4 A/mm, and a gate leakage current, IG, below 5 nA in a gate voltage range VG=+10 to −10 V.
FIG. 6A shows measured (solid lines) and simulated (dashed lines) transmission-frequency dependencies of the isolation and insertion losses for the integrated RF switch, with a simulated isolation of a single element switch for comparison. FIG. 6A illustrates that use of the multi-element π-layout with very small parasitic parameters achieved by the monolithic integration results in a significant improvement of the device performance at high frequencies. Additionally, the modeled and measured parameters closely agree. The switching powers of the switch MMIC are nearly the same as for a single series MOSHFET, i.e., in excess of 30 W/mm2. FIG. 6B shows a transmission-power dependence for an illustrative MOSHFET RF switch. As shown, an increase of the input power at 10 GHz up to 5 W (limited by the available power source) showed no noticeable change in the insertion loss or in the isolation. The level of the third harmonic distortion was below −40 dB.
While the above-referenced MOSHFETs perform well as high-power RF frequency switches, some limitations make them difficult to use. For example, the source/drain/gate contacts are ohmic contacts, which require annealing to obtain the desirable performance attributes. The annealing adds complexity and cost to the generation of the devices. Additionally, the three ohmic contacts occupy valuable areas on the surface of the chip and may adversely affect the reliability of the device.
In view of the foregoing, a need exists to overcome one or more of the deficiencies in the related art.